EPA Imidicloprid Review Underway
We take issue with Field acceptable levels that the EPA proposed in their Preliminary Draft. This is our comment to the EPA
Our comment letter to EPA follows below.
Environmental Protection Agency Docket Center
1200 Pennsylvania Ave. NW.
Washington, DC 20460-0001
Re: Pollinator Ecological Risk Assessments: Imidacloprid Registration Review
To whom it may concern,
We would like to offer some comments on the regulatory endpoints that are proposed in the first Preliminary Pollinator Assessment to Support the Registration Review of Imidacloprid (January 4th, 2016).
Below is a figure from our paper (Rondeau et al. 2014), with the toxicity endpoints and the colony-level feeding exposure endpoint shown along with our summary of time dependent toxicity data for imidacloprid.
Indeed, the EPA has chosen the 10-day LOAEC and 48-h LD50 to be safely on the low side of most of the reported results in the literature. However, we feel that the colony level exposure of 25µg/L as determined from the Tier 2 study results should be viewed with much caution. The recommendation of <25µLa.i./L is shown as the red vertical line on the plot. Nectar consumed at 25µLa.i./L with typical daily consumption of 20µL/day yields a daily dose of 0.5ng a.i. per individual. A single day’s exposure, 0.5ng/bee, exceeds by a factor of 2 EPA’s accepted LOAEC of 0.24ng/bee. It is very hard to imagine that the cumulative dose to individual bees at this level of contamination would not give rise to neuro-toxic symptoms.
Curiously, the 25µLa.i./L exposure rate corresponds on our graph to the LT50 time of about 20 days, roughly the age when nurse workers enter the field foraging force. With the colony getting much of its nourishment from the contaminated nectar, there may be little need for a robust work force aged much more than that. Clearly the colony is providing services that transcend the health of individual bees. In fact, it can be expected that there may be colony-wide paradoxical effects. The colony response to a shorter lifespan of individual bees could very well be acceleration of brood production. If colony health was measured by the amount of capped brood, such a seemingly positive effect would be interpreted incorrectly.
In our study we found that the results from many diverse toxicity studies for imidacloprid on insects could be understood in terms of a simple power law scaling,
LT50 ∝ D tP
Where D is the dose rate, t is time and P is the power law exponent. We discovered that much of the honeybee toxicity experiments can be unified with such a scaling law where the time exponent is approximately 2.
Although there is an abundance of toxicity data for imidacloprid and honeybees, there is still missing a good study that can determine the time when such t2 scaling is likely to come to an end. Absent such a study, the prudent policy is to assume that the scaling we present continues indefinitely until individual bees die. If you use bees as proxies for solitary pollinators and other beneficial insects, then you are left without the benefit of robust colony services to maintain health in the presence of pesticide residue. Instead the best estimate we would have would be individual toxicity scaling, with a maximum time based upon natural insect lifetimes or reproductive cycles. This will vary dramatically from species to species, and could approach good fractions of a year for insects on an annual cycle. Unfortunately, the regulatory process is not as simple as assigning a threshold guideline concentration for which to stay below, since the organism life span becomes part of the equation.
The EPA’s continued adherence to the concept of threshold toxicity levels should be questioned with pesticides like imidacloprid that are designed to bind strongly to synaptic receptors and which act to directly stimulate the post synaptic junction. Relatively few molecules of the pesticide can have long lasting effects, as we demonstrated with a model in our paper. This is in contrast to acetylcholine esterase inhibitors, where the action of a just few molecules does almost nothing, since there would still be many acetylcholine esterase sites to clear the junction of neurotransmitter. Hence, for this latter class of chemicals (organophosphates), there is a natural threshold built into the toxic effect, namely the point where concentration of pesticide is sufficient to inhibit a large fraction of the acetylcholine esterase sites.
EPA’s history of successful regulation of the organophosphate insecticide can be seen as a vindication of the threshold theory for the acetylcholine esterase inhibitors. The neonicotinoids present the EPA with a major change in mode of action. Hence a fundamental change in the regulatory framework that backs away from the threshold concept and looks more deeply at the accumulation and persistence of toxic effects is required (Sánchez-Bayo & Tennekes 2015).
Gary Rondeau, Applied Scientific Instrumentation, Oregon, USA
Francisco Sánchez-Bayo, The University of Sydney, NSW, Australia
Henk A. Tennekes, Experimental Toxicology Services (ETS) Nederland BV, Zutphen, The Netherlands
Axel Decourtye, ITSAP-Institut de l’Abeille, Avignon, France
Ricardo Ramírez-Romero, Universidad de Guadalajara, Jalisco, Mexico
Nicolas Desneux, French National Institute for Agricultural Research (INRA), Sophia-Antipolis, France
Rondeau, G. et al. Delayed and time-cumulative toxicity of imidacloprid in bees, ants and termite. Sci. Rep. 4, 5566, doi:10.1038/srep05566 (2014).
Sánchez-Bayo, F. & Tennekes, H. A. in Toxicity and Hazard of Agrochemicals (ed Marcelo L. Larramendy) Ch. 1, 1-37 (InTech Open Science, 2015).
Once I had the RTL-SDR radio working as a pan-adapter for my IC-751A, I quickly became interested in seeing what I could do with all of those signals that were apparent on the SDR radio’s waterfall. My first goal was to see if I could receive the complete JT65 and JT9 signal band, with both modes usually in a band region up to 4 kHz wide on the digital sections of the main ham bands. The WSJT-X program is happy to decode signals with this audio bandwidth, but the native IC-751A only has about 2.8kHz audio bandwidth on the upper side band mode. So this could be a simple enhancement of the existing receiver if it really worked. I set up the RTL-SDR and various software components as described previously. But first I wanted to know just how good is the SDR receiver really?
There is a big advantage to have the RTL-SDR on the IF of a good multi-band receiver. By the time the RF signals get to the 1st IF mixer, they have already been filtered by the low-pass band filters of the parent receiver, which effectively eliminates interference from local high power FM stations that can degrade the performance of the RTL-SDR. Hence, the RTL-SDR should be able to perform just about as well as it ever will in this situation.
Questions immediately arise when trying to set the proper receiver gain. There are many places to turn up or down gain levels. Here is a list starting at the antenna:
- The antenna tuner can dramatically increase the RF signal level when the antenna is resonant.
- The IC-751A has an RF preamp before the 1st mixer that can be deployed.
- The RTL-SDR control allows “Tuner AGC” or fixed Tuner Gain from 0 to +50 dB.
- The RTL-SDR also allows “RTL AGC.”
- The HDSDR software has AGC and Volume controls that affect the demodulated output levels.
- The sound card channels have level adjusters.
- WSJT-X has an input audio gain setting slider.
What I discovered was that the RTL-SDR receiver needed as much RF gain as I could give it to get comparable results to the IC-751A. A tuned antenna helped the RTL-SDR much more than it helped the IC-751A for clean reception. With the antenna well tuned, the IC-751A RF preamp was less important than if the antenna was not resonant. The RTL-SDR Tuner Gain needed to be all the way up, +50 dB, for best performance. “Tuner AGC” seemed to accomplish the same thing as just setting the gain at 50 dB. Once past the Tuner Gain setting, all of the later gain adjustments for the demodulated audio had little effect unless you turned things down completely. I followed the WSJT-X recommendation to set audio levels such that the quite band noise level showed about 30 dB on the WSJT-X level indicator.
The figure above shows what I manage to receive using the two receivers on the same signals. From the antenna to the 1st IF stage of the IC-751A, the signals travel the same path. After that, the RTL-SDR dongle’s tuner and 8 bit ADC generate digital signals that HDSDR converts to an audio stream, which is then sent on to one instance of WSJT-X. The other WSJT-X window is decoding signals processed by the IC-751A in the conventional way. Hence, just as I did before, we have a simple method to compare the relative receiver performance using the reported JT signal to noise reports generated by the WSJT-X software as the metric. One thing you can immediately notice is that the HDSDR bandwidth is large enough to cover both the JT65 and JT9 signal regions of the band whereas the 751A’s filters limit the bandwidth to about 2.8kHz. The other thing you might notice if you look carefully, is that some of the splatter that you see on the IC-751A in the presence of a very strong signal is not present on the SDR radio. Notice, at time 19:46, the strong signal at ~1200Hz, and the splatter 2200 – 2500 that is absent on the upper image. Goes to show that arithmetic doesn’t generate inter-modulation images, where as real mixers and amplifiers can.
So on to the noise performance data summary from thousands of JT65 and JT9 signal reports, generated by simultaneously decoding the signals on the two radios. The table below summarizes the experimental results. The first column is how much better the IC-751A is compared to the the dongle using HDSDR.
With the non-resonant antenna, especially on 40m, the RF signals are weak and need amplification to achieve the same performance as the IC-751A. However, you can see once the RTL-SDR dongle is given enough gain by tuning the antenna or using the preamp, then there is really very little difference in the quality of the decoded signals that WSJT-X provides between the software and hardware radios.
At first I found this quite surprising, bearing in mind that the RTL dongle has merely an 8-bit ADC. Realize that the range of signal levels the WSJT-X can decode, and I accept for comparison, is about 20 dB for JT65 and >40 dB for JT9 signals. Now bear in mind that with 8 bits, the RTL’s ADC will be limited to a maximum of 48 dB dynamic range. So, I guess this “just fits” more or less. Very seldom to we bump into the top end of the dynamic range of the JT signals. Rather we are almost always struggling to pull them out of the noise. So as long as there is sufficient gain at the front end, the RTL tuner can do its job. It begs the question about any advantage an SDR with a higher resolution ADC might have. Maybe someday I’ll get one and I can find out!
There is little doubt that the RTL-SDR tuner dongle radio receivers are the hottest new thing for the amateur radio experimenter. Re-purposing the $10 tuner dongle to be an extraordinarily wide-band software defined radio is the subject of countless internet articles and videos. Here I am adding my experiences to the crowd. What I will describe are the steps required to use one of these devices as a full-band pan-adapter for a conventional receiver. Once that is working, then we can look more closely at the actual RTL-SDR receiver performance, compared with the native performance of the parent receiver. I’ll discuss the performance of RTL-SDR receiver in another segment. For now, let us get this up and running.
First a diagram of what we are about to do:
My RTL-SDR is the common RTL2832U chip with R820T tuner. The device can tune from about 30 to 1700 MHz. To be useful for the HF ham bands, some up-conversion is required. Many HF receivers employ a first IF stage, mixing upward to a 45 to 75 MHz intermediate frequency. This is perfect for the RTL-SDR tuner. Tapping into the IF after the first mixer is easy on the IC-751A since there is an unused connector on the RF board that is designed for scope monitoring of signals at this location. On my old IC-745 I had to solder in a resistor at the right spot to get the signal. Usually a little study of the receiver schematic will make clear where you need to tap into the 1st IF signal, and whether the radio design will make this easy or not. It is important to tap in right after the mixer and before the 1st IF band-pass filter, since otherwise the pass band that you can tune with the RTL-SDR as a pan-adapter will be severely curtailed by the IF filter. Ideally, use a short piece of 75 Ω coax to match to the RTL-SDR tuner input impedance (but 50 Ω is okay too).
Now we can bring up the RTL-SDR in the HDSDR software. If you are doing this for the first time, some good instructions are located here. Of the various SDR radio programs that are floating around the internet, HDSDR seems to be the best for interfacing the RTL-SDR with another radio and then being able to co-ordinate the tuning of the two systems. When you first bring the dongle up in HDSDR you should be able to see HF band signals when you tune HDSDR to the IF frequency. This makes it confusing to remember where you are tuned, so HDSDR has the ability to shift the tuning display by the IF frequency offset (RF front-end frequency options & Calibration tab). I suggest tuning the receiver to a well-known strong AM station (WWV at 5.0 or 10.0 MHz is a good choice) and then playing with the IF-offset until you have HDSDR tuned correctly. Including a global offset of 10kHz is a good idea so that the IF zero beat is not in the middle to the HDSDR’s audio output band. Additionally, it is important to make sure that the side bands are not mirrored, and the corrections for tuning USB, LSB, CW, etc. match that of the parent receiver. Tuning JT65 signals can help clarify this to make sure they are not mirrored (and un-decodable!). For whatever reason, I needed to check the “Swap I and Q Channel for RX Input” to achieve the correct result. I also set a USB offset of -2940 Hz so that HDSDR and the 751A would tune together in USB mode.
HDSDR will be the master of all, so we set up the HDSDR Omni-Rig interface to talk to the IC-751A via the serial RS-232 port. At this point, it should be possible to tune with either HDSDR or with the receiver directly and have them follow each other. As a pan-adapter, you are ready to go. You can open the band in HDSDR and see signals across the band. A simple click and you will tune to that signal on the 751A. For phone operation, you are done – have fun.
With everything working, the pan-adapter begins to provide useful information. The example snapshot above shows a smattering of signals on the USB phone section of the 20 meter band where the HDSDR is tuned. It appears that the IF pass band of the 751A drops the signal level on the RF waterfall a little, so you see less background in the center of the waterfall where the parent receiver is tuned. With the pan-adapter synchronizations working, clicking on a signal will immediately return the receiver so you can copy the signal. In fact, you can hear the audio on either the parent receiver, or on the computer speaker from the HDSDR demodulated audio output.
For digital modes, we need to run other programs to decode the digital signals. WSJT-X for JT65 and JT9 signals and FLDIGI for a multitude of digital mode transmissions are the programs I use most often. Besides running those programs concurrently with HDSDR, we need to plumb in the audio and control signals to these applications. For most straightforward operation, the digital mode program will be connected to the audio channels of the tranceiver just as you would normally for the stand-alone configuration, but the rig control must now come from HDSDR. To do this you need a Virtual Serial Port pair. There are a couple of possible programs that are available to do this for you. I’m using VSPM from Steve Nance, K5FR. You will have to write to him with your call sign to get a copy for you to use. Reading between the lines, it looks like Steve’s program, updated and expanded, is being sold as Eltima’s Virtual Serial Port Driver, which I’m sure also works, but is not free. A second free program, Eterlogic VSPE, also might work for you. VSPE has some nice additional features, like the ability to have port “splitters” as well as port pairs, but I also experienced some BSOD (Blue Screen Of Death) errors when using this program on my old Windows Vista laptop. Whatever you use, once you install a port pair you can now connect the HDSDR’s “CAT to HDSDR” port to the rig control communication port in WSJT-X or FLDIGI. HDSDR talks in “Kenwood TS-50S” language, so the digital rig interface should be set up accordingly.
So far, the RTL-SDR dongle is just showing you a wide view of the ham band and the parent transceiver is operating in the normal way. But there is more that we can do with the RTL-SDR dongle than just look at the big picture. If we pipe the output of HDSDR’s demodulated audio into a digital mode program, we can use the RTL-SDR dongle as the real receiver rather than just the panoramic view generator. To do this we need another little program called Virtual Audio Cable. This program is not free, but is inexpensive for what it does and how well it works. Once installed, the program can generate pairs of sound-device ports, virtual audio cables, that can be used to pipe audio streams between applications.
We can connect the “RX Output” from HDSDR into a virtual cable, then connect the other end of the virtual cable to, for example, the WSJT-X sound card input so that we could decode JT65 and JT9 signals directly from the RTL-SDR dongle. Once you discover the flexibility of Virtual Audio Cables and Virtual Serial Ports you will come up with many interesting configurations of software components that can be strung together in interesting ways.
I found that the RTL-SDR dongle can be used as a serious receiver of digital-mode signals. However it lacks any transmit capability so you must return to the parent transceiver for that function. Beware that the frequency calibration “going around the loop” between the RTL-SDR receiver and the conventional transmitter must be carefully maintained. I would fine tune with the USB offset number in the RF calibration tab. The wide audio bandwidth that the RTL-SDR can generate is not present in the conventional transmitter. Just because you can decode a JT9 signal up at 5 kHz on the audio of the RTL-SDR does not mean that you can point the transmitter there and have it work! You can use some of the same principles, using VSPs and VACs to connect other SDR radio programs together with a slew of audio processing and decoding programs. The possibilities are almost endless — you better go get one of these and start playing with it!
True understanding of a problem is confirmed when one can validate theoretical model predictions with measurements. It is too easy to believe pretty pictures that modeling programs produce, especially when making meaningful measurements is so difficult. In this article I try to take the bull by the horns and see how close I can come to declaring that I understand my antennas.
Last time we discovered that the noise performance of my two old radios was essentially identical to one another. This means that I can use the two receivers to listen to signals from my two antennas and be able to make comparisons between them. When digital mode JT signals are decoded, the software determines a signal-to-noise measurement for the received signal. We can use these measurements which we get for free with the WSJT-X software for every received signal to quantitatively make comparisons between the two antennas. In practice this means leaving the two receivers tuned to the same JT65 and JT9 band, have both receivers running an instance WSJT-X software, and recording all of the transmissions received by both radios.
In my case, I have a 40m loop that is about 30 feet high and an off-center-fed dipole (OCFD) at about 75 feet up in my tall Douglas fir trees. Both of these antennas are horizontally polarized and have distinct directional properties that are unique to each geometry. This means that I would expect signals from stations a various locations to be preferred by one antenna or another.
To begin with we will look at the predicted propagation pattern of the two antennas on the 20m band. Since I am now getting down to making comparisons for real antennas with a model, I spent some time adding in as many parasitic elements into my NEC model as I could, and included both of the antenna geometries in the same model – just substituting the driving point of the unused antenna with the matched transmission line impedance. Besides the two antennas I also included the aluminium rain gutters on the house in the model, and I did my best to correctly locate the antenna elements as they really are laid out. The NEC model and the two radiation patters for the two antennas are shown in the following figures.
The pattern for the 40m loop antenna is less complex at 20m, since there are fewer harmonic resonances on the loop.
Since the OCFD is considerably higher than the 40m loop, it generally has better low elevation gain; but the two antenna patterns are really quite difficult to directly compare to one another. Fortunately, 4NEC2 can give us tabulated data for both patterns. Since we will be looking at differences in signal levels, we can also look at differences in the predicted model radiation patterns. I don’t really have the tools to do this with the full 3D pattern, so instead I just looked at the difference in total far-field radiation at the 1o and 15 degree elevation ranges where long distance propagation is most often successful.
With the model results now in hand, let us look at the results of the JT65 & JT9 signal comparisons. After collecting a few days worth of JT signals and spending some time pouring over the Excel spread sheet I was able to generate the antenna difference radiation pattern. Rather than having a nice uniform distribution of signals, there are large concentrations of data points from US, Japanese and European hams and otherwise a fairly sparse azimuthal distribution.
Is there agreement? Hard to say, but some things are clear. Usually the OCFD does better than the Loop, often by 5 dB or more. We see various “lobes,” although they do not necessarily line up very well with the ones predicted by the NEC code. The lobe aimed at the EU at 30° seems in place, and stations due North are suppressed on the OCFD as we would expect along the wire direction. The polar plotting is best for comparison with the NEC model. Below we show the same data as function of compass bearing angle along with the a measure of the statistical errors present in the data, and some geographical reference points.
The graphs above shows the result after quite a bit of data manipulation in Excel. Remember that peaks in the above chart can come about because of either a strong transmission lobe for the OCFD antenna or because of a null on the Loop antenna. Many factors can give rise to a spread in the gain difference at a particular bearing angle. The actual propagation path for a particular transmission might be at any of a range of propagation elevation angles, all of which are it different for each antenna. The transmitters have unknown polarization and the antennas will respond differently according to the actual polarization vector. The radiation patterns I’ve presented are just the total gain and do not consider this added subtlety. Nevertheless, there are definitely directions that favor one antenna or the other.
The data file is included here as an example and template for others to use. Excel Antenna Comparison Calculation Template The template file has some useful Excel formulas embedded in the worksheets that will calculate Latitude and Longitude from the Grid location code, and will then calculate bearing and distance from your location.
Several steps are required to get the cleanest results.
- Only signals received simultaneously by both receivers are considered.
- JT65 signals when both receivers showed signal levels less than -5 dB s/n are included. Strong JT65 signals are rejected from the data considered.
- Grid data is added to signals where it is possible to unambiguously determine the grid.
- All signals from a particular grid square are averaged to get a single number of each grid square.
Once the averaged signal differences for each grid point are determined, this data is further processed .
- Latitude and longitude are determined from the grid location code.
- Bearing and distance are determined from the latitude and longitude of the transmitter and receiver locations.
- Data for distances less than 500 km is excluded.
- Signal differences are plotted on a polar graph.
The details I presented above for the 20m band show the level of data analysis possible with the kind of information that is relatively easy to collect in just a few hours of listening with a couple of radios. Even a much more cursory look at the data is valuable, however. Merely taking averages for all matching transmissions on a given band can give a single-number figure of merit for comparison of two antennas.
Here is such data for my two antennas for the four bands where the loop works well.
First thing to notice is that although the OCFD up at 75 feet does significantly better on the lower bands, 40, 20 and 15 meters, the Loop wins out on the 10 meter band. This was a bit of a surprise, but goes to show that height is not always your friend.
Also notice the rather large standard deviation. In some sense, this is just a measure of the depths of the lobes of the two antennas.
Direct comparison to the NEC models was marginally satisfactory. However, there were many small uncertainties for both of my antenna geometries which, when all combined, proved to make the direct comparison with the theory less than perfect. The models are only as good as the input data, but the radio signals tell no lies.
In the future, doing a similar experiment with a vertical antenna as a reference might be a good plan. The uniform radiation pattern from the vertical would provide a constant reference that would allow the pattern of the test antenna to be more accurately determined.
I entered second heaven when I got my second radio. My original 32-year-old Icom IC-745 got placed on the back shelf when the new 29-year-old Icom IC-751A arrived. As much as I enjoyed the old IC-745, I was after a little computer control, and the 745 predates the idea that people use computers. The 751A, just a few years later at the dawn of the PC age, at least had an optional serial computer interface.
Once I had two radios I could not help but to start to make comparisons. Both radios do an admirable job, but I began to notice some small differences in the way they would handle strong narrow-band digital signals. When I used the IC-745, I would turn the AGC off and manually set the RF gain so that I was just picking weak signals out of the static noise floor. This meant that strong signals would be over saturated, but the 745 didn’t mind that too much. When I tried to use the IC-751A the same way, the strong signals would generate signficant amounts of in-band spurious interference, mostly third-order intermodulation products (IMD3).
What happens if you don’t have your levels set right? Most hams know that if you overdrive PSK signals during transmission, there can be IMD3 spurious emissions that can disrupt the band. This appears on PSK signals as multiple “railroad tracks” that spreads out over the band. Less well-known are the effects that come from improper receiver level adjustments. The tricky part is that we need to process a very large dynamic range of input signals, all right next to one another. The IMD3 problem shows up on receive as well as on transmit. Here the problem is the mixing of harmonics of two in-band signals, where the mixing products appear back in the pass band. The “third order” products are generated by the mixing of the fundamental frequency one signal with a second-harmonic of the second and vise versa. With frequencies A & B, the in-band mixing products will be 2A-B and 2B-A. If A-B = Δ, then you will find spurs at A+ Δ and B – Δ.
The picture above shows a clear example of receiver IMD3 distortion. The IC-751A had the AGC turned off so when the strong signal in the center of the waterfall began broadcasting, the front-end stages of the receiver were overdriven. The same signals in the IC-745, with the AGC on, don’t have the problem. Note the strong “mirror” effect of spurs across the strong signal. If you were operating this way, you may be tempted to latch onto one of the traces on the right side ( above the IC-751A label in the figure). The nature of PSK is that these spurs are decodable, but if you try broadcasting there, the sender will not be waiting for you.
Before we move on to the measurements, lets look at another set of spurs that have a completely different origin.
Note the two spurs at 1600 and 2680 Hz on the IC-745 waterfall above. There is another strong signal in the picture, but the spurs don’t appear related to the frequency difference between them. In this case we are looking at audio harmonics, the third and fifth, of the audio signal at about 550 Hz. The odd harmonics will appear in the audio if you start clipping it. In this case, the problem was that the sound card was being driven too hard on that strong signal. The solution was simply to reduce the input level to the sound card.
It took me a while to realize that the IC-751A had significantly more RF gain than the IC-745. If I set the manual RF gain on the 751A at about 60% of full, I had similar gain to the IC-745 at full. But before I figured all of this out, I thought that the IC-745 had better noise performance the IC-751A. But how to know for sure?
My test consists of splitting the antenna signal with a 50 Ω power tee so that each receiver gets the same signals from my antenna. Then I set up both receivers to have almost identical noise characteristics on the FLDIGI waterfall. I run two copies of both FLDIGI and WSJT-X during the comparison. Running WSJT-X, I set the input levels to have about 30 dB noise level between transmissions on both receivers. This gives the decoders plenty of dynamic range and again matches the overall gain of both receivers for the digitized signals. Now we just listen for a while and accumulate JT65 and JT9 signal reception reports from both receivers. If everything is identical, you should get identical signal-to-noise reports on both rigs. Comparing the two waterfalls can show where there are problems with one rig or the other.
The JT65 and JT9 signals on a busy Saturday afternoon shown in the waterfalls above illustrate the dilemma. Many of the transmissions are very strong, and weak signals may be intimately interspersed among the strong signals. I found that I could get significantly better noise performance on a crowded band by reducing the RF gain below the point that the AGC would normally operate, especially on the 751A. The waterfalls above were recorded with only 40% of max RF gain on the 751A and 75% of max gain on the 745. Under these conditions, the signal reports back from the WSJT-X program are virtually identical between the two radios. Both of my receivers can be made to show strong IMD3 spurs if the front end gain is allowed to get too high. The AGC helps in this regards, especially with the IC-751A because this receiver has quite a bit more gain head-room in the RF amplifier than the IC-745. Under extreme conditions manually turning the RF gain down even further seems to help. JT65 signals can become a speckled smear across the band when there is IMD3 distortion from multiple strong signals.
In the midst of this exercise, my neighbor three blocks aways started calling on the PSK band just below the JT band in frequency. This caused some trouble for the IC-745. My neighbor’s +60 dB over S9 signals, only 6 kHz away from where I was tuned, made a mess of the strong JT signals. I found it curious that the problem was greatest with the strong signals. Clearly this is one place where the IC-751A had superior performance.
Measuring accurately the comparative noise performance of the two radios required analysis of the reception logs created by the WSJT-X programs. The reporting on JT65 signals is compress at the high-end, the program never sending more than -1 dB signal level report. JT9 on the other hand, robustly covers signal-to-noise levels from +15 dB to -28 dB. This makes the JT9 signals much better beacons for our tests, however there tend to be more JT65 signals to be received. I looked at both modes, but for JT65 I don’t consider strong signal reports larger than -5 dB in the receiver comparisons. In short order it is possible to collect several hundred report pairs that can be used to get good statistics. You can use my RX_compare spread sheet as a template to do something similar if you wish.
It is not too surprising that my two radios have essentially equivalent noise performance. They were both considered “hot” receivers during their day. I would be really curious to repeat this little test, comparing a modern direct conversion SDR receiver with the classic superhetrodyne architecture my receivers have. That might show some real differences because of better IMD3 suppression in radios without so many mixers.
Now that I am convinced I can adjust the two receivers to accurately reflect the antenna noise performance without introducing their own bias, I can proceed to compare my two antennas. Stay tuned!
There are a few ways to build multi-band HF antennas. These include multiple resonant dipole arrays, trapped dipoles, resonant loops and the off-center fed dipole, OCFD. I’m going to look at the OCFD because it is relatively simple to construct, yet gets quite decent performance. I also am fortunate to have some nice tall trees from which to hang such an antenna.
- Simple construction
- Multi-band operation
- Horizontal polarization
- Low radiation angle with good efficiency
- Has preferred propagation direction which can be aimed
- Usually requires an antenna tuner
- Usually requires a matching balun
- Has lobes and nulls which may limit coverage to some areas
- Is not a beam
The ham bands are laid out across the radio spectrum at rational frequency multiples of each other – more or less. This wise decision, made many years ago, allows relative ease at using the same piece of wire to receive more than one HF band.
The chart above plots the standing wave amplitude of waves on a wire that all start with zero amplitude on the left side. If you cut the wire where you again have an amplitude zero, a standing wave would be supported on the wire. Arbitrarily, I chose to express length as the phase on a 40m long wire. Note that many of the bands are close to resonant at the 360° (40m) point. The exceptions are the red crosses at 360°, namely the 160m, 60m, and 30m bands. An antenna made this long will completely fail with the 160m and 60m bands, and will have some difficulty on 30m. In fact, 30m is problematic in general because the actual ham band is not a very close multiple of the others, so will often require more of a compromise. The lowest resonant band on the 40m wire is the 80m band, hence a wire this long is classified as an 80m dipole. If you didn’t have enough space for the 80m antenna, you could consider stopping at the 180° point on the chart, where again there is another convergence of zero amplitudes for the 40, 20, 15, and 10 meter bands; or maybe you could stop at 250° and attempt to capture the 60, 15, 30, and 12 meter bands. Picking the zero convergence selects the overall length of the wire, but we still have to pick the dividing point where we drive it.
There is a nice design that picks the point at 120° as the driven point (the small black circle on the plot). The 80, 40, 20, 17, 12, and 10 meter bands all have roughly the same amplitude at this phase along the wire. This means that they can all be driven with a similar amount of effort from the transmission line (about the same impedance for all bands). There are two bands that are not driven very well at this point, the 30m and 15m bands (black x in figure). A commercial antenna system, the Buckmaster 7-band OCF Dipole, exploits this property. The big advantage of this design is that you can make an antenna that does not require an antenna tuner by utilizing the constant impedance of the happy convergence that happens at 120 degrees.
I didn’t go for that design, however, because I really want the 15m and 30m bands as well. I felt that both of these bands are more important than the 12m band, which is poorly driven with my design. I picked about 103° along the 40m wire as the operating point for the feed. The red oval shows the wave amplitudes for the various bands at my chosen feed point. There is quite a range over which I need to drive the antenna. Hence we will likely find that it is not possible to get a fixed impedance for all bands, but will instead need to provide an antenna tuner to aid with the matching. At this point I broke out the computer simulations for fine-tuning the lengths, to see what the radiations pattens should look like, and to get a better idea of how I would have to drive the antenna.
I had tall support points in the trees for the feed point and the far end of the long wire. The short wire would be pulled down to a lower fastening point. The model, shown in the figure on the right, includes the feed coax. The feed is not connected to either main element since a balun is used to drive the dipole. However, it is an easy matter to connect the vertical feed line to one side or the other and add a choke somewhere on the feed to get yet more interesting resonances. Such tricks are employed with “Carolina Windom” antennas. My model runs suggests only minimal changes if you connect up a vertical element so we will not further consider this modification here. As-is, the vertical cable run is not particularly resonant with any band and has little effect whether it is included or not.
Specifications for the AF7NX OCFD:
Long leg: 93′ #14 THHN copper wire.
Short Leg: 37′ #14 THHN copper wire.
Height of feed point: 75′
Angle of short leg: 40° from horizontal (flatter is better)
Feed: 4:1 Balun to 75 Ohm CATV cable to Tuner.
The figure below shows the antenna tuning plots. (Click on the plot to get the big picture.)
The antenna is at a resonance when the reactance term (red line) crosses zero. Those points should be close to the ham bands a 3.5, 7, 10.1, 14, 18.1, 21, 24, and 28 MHz. Most resonant frequencies are close to the desired ones except for 24 MHz (12 meters) which we knew was not going to be possible. At the resonances, the resistive term will be the antenna driving impedance. There is quite a range over which the driving impedance varies, from ~90 Ω at 7 MHz to ~700 Ω at 10.1 MHz. The best we can to is select something in between 90 and 700 ohms. With my 75 Ω co-ax using a 4:1 balun brings the impedance of the driver to about 300 ohms, which is a good compromise.
You can see the band resonances on the SWR-300 Ω plot. Portions of the 80m, 20m, 15m, and 10m bands could be used without a tuner if you have a 75 Ω transmitter, but our plan is to match with an antenna tuner at the transmitter anyway. The antenna tuner will ensure that reflected power returning from the antenna will not be sent along to the 50 Ω transmitter. Hence, any power reflected from the antenna must be re-reflected back to the antenna again at the tuner. If there were no resistive losses in the feed components, we wouldn’t care at all about the SWR, since eventually all of the power would be radiated from the antenna after several bounces back and forth to the tuner. But there are fintie losses in the cable, so too many reflections will sap transmitter power. A good discussion of line losses and SWR can be found in this 2006 QST article. The numbers in the brown shaded column in the table below come are derived from a graph in that article. The table also shows some of the numbers from the NEC model runs (shaded blue) for each ham band.
For our 80 foot run of Belden 9118 coax, the overall transmission efficiency is respectable for all but the 12m band, where we expected problems; even so, 12m is marginally useable. The 30m band is also a little less well-coupled than one might hope, but this is a particularly difficult band to get to resonate well with the other frequencies on the wire; 17m has the same problem to a lesser extent.
It is worth pausing to consider the effects of the SWR mismatch on both the transmit and receive operation of the antenna. It doesn’t matter which way the signals are going, there will be losses associated with the SWR in the transmission line, but the overall effect on performance is quite different in the two cases. On reception, the critical figure of merit is the signal to noise ratio at the receiver and NOT the total signal level. Almost always, the limit to noise performance is “atmospheric” band noise and not the noise limit of the receiver. Lets consider the 12m band where we have 5.3dB of line loss, mostly associated with SWR from the poor match. If we are listening to a transmitter coming toward one of the high-gain lobes of the antenna, we can expect that the signals on the antenna will be ~7.8 dB larger than the noise compared to what would come from an isotropic antenna. On the way to the receiver we will lose 5.3dB, but atmospheric noise received by the antenna will also be attenuated by the same amount. If the receiver has enough clean gain to make up for this loss, we might be quite happy with the antenna performance during reception. On transmit, however, all we care about is the relative amount of power aimed at our target receiver. The 5.3dB line loss means that less than a third of the power from the transmitter will ever get out; two-thirds turned into heat in the transmission line. If you normally run your digital mode at 15W now you will have to use 45W. Of coarse, the antenna gain pattern still matters to be able to throw the power where you want it.
The table lists the maximum antenna gain for the low-elevation lobes. Let us look at the radiation patterns that NEC2 gives us. Plots are the elevation and horizontal patterns for six of the bands supported on the antenna.
Notice the progressively lower maximum elevation peak as you progress to the higher frequency bands. Looking at the horizontal pattern notice the low gain off of the end of the antenna in the Y direction. Also note that the maximum for the radiation pattern is not strictly broadside to the antenna except for the lowest fundamental 80m band. For the higher bands the largest lobe swings around to pointing 50 to 20 degrees from the antenna wire, depending on the band. I’ve aimed the antenna so that Europe is about 35 degrees from the direction the wire is pointing to attempt to take advantage of one of the high-gain lobes.
Finally, one last pretty picture of the propagation “rose” for the 20m band. The pattern is complicated with multiple lobes at several elevations. The nice strong low elevation lobes are quite narrow and oddly spaced so it is hard to know exactly what the antenna is actually pointing at. The strongest stations you hear might just happen to be lined up to the pattern, and you might miss stations seemingly close by that are just outside the best directions. This antenna just adds a little more contingency to the already very contingent whims of propagation, but it is simple and surprisingly effective.
I’ve got this antenna up in the air now. It works pretty well. In a future post we will look more carefully at its actual performance.
Broadcasting’s use of abbreviations and obscure notation started with the original radio-men, when every character required some pain to transmit. Today’s tweeters and texters have invented their own slang for much the same reason, but without the long linguistic history that accompanies radio lingo. When you first enter the ham world, you have to cross the language barrier, even if it isn’t particularly high. You turn on the radio and connect to the digital stations and immediately see CQ CQ CQ DE K7BT K7BT PSE K streak across the screen, highlighted in red. What can that mean? Here is a translation and lexicography of some of the more interesting bits of ham lingo in no particular order, but starting with the message in red.
CQ – “Calling any station” comes about from the sounds of the letters C and Q spoken in French, and meaning “sécurité”, here translated roughly as “attention.” This is the universal “all call.” When my digital radio program sees CQ it turns the line red to alert me to the call.
DE – “from” from before English was the lingua franca, often followed by the sender’s call sign as in our message above. Usually the call sign is repeated more than once for clarity. This universal calling convention, along with call-sign naming conventions, allow automated listening programs to pick out a station’s origin from the jumble of text being received. The very useful program PSK Reporter utilizes this convention of ham lingo to amazing effect.
K7BT – a call sign, one of millions, the unique moniker of every amateur operator. The characters before the numeral are unique to the country granting the call sign.
PSE – “please” when every letter costs a good fraction of a second, you abbreviate everything.
K – “done transmitting – your turn – go” This is a curious one, since I can find barely a reference to it anywhere in the simple form of ‘K’. It seems to be so ingrained as a form of “over” that it goes without saying in the ham lingo guides. Its more formal cousin, “SK,” derives from the obsolete American Morse Code sound-alike for the Western Union code “30” which means “No More (end).” Shortening the very final “SK” to simply “K” seems to imply that this is just the end for now, your turn.
SK – “The End, No More, All Done” when the receiver should not expect another response from you. Derived as described above, but this term also has another meaning. SK is also short for “Silent Key,” meaning an operator who has passed away. After all, the “silent key” is indeed “no more – all done.”
73 – “best regards” is another of the Western Union telegraph codes that migrated into the ham world. Not to be confused with its cousin…
88 – “love and kisses” – Western Union again. This is would be a nice way to end a message to your YL or XYL.
YL – “young lady” or just about any woman, young or old, unless specifically your XYL.
XYL – “wife” as in ex-YL. If you think that too denigrating, the guys don’t have it much better.
OM – “old man” is just about any operator that is not a YL. If you found that funny, you could laugh like a ham.
Hi Hi – “ha ha” apparently because in the dits in the morse code “hi hi” translates into “he he he he he he he he he he he he” which sort of sounds like laughter. Which gets us to the confusing ones…
FB – “fine business” — Really? who says that? Everyone if you are a ham. But I think it translates better into “fabulous” or maybe just “ok.” Ambiguity continues with…
TU – “thank you” obviously, but less obvious when …
BTU – “back to you” is another form of “over.” So it seems like TU should just be “to you” as in sending “30W TU.” Am I thanking him or am I sending him my transmitting power – or both?
TEST – “contest” because one of the major pastimes of amateur radio operators is spent “contesting” for more contacts. It took me a while to realize that the ham sending CQ TEST wasn’t after help to fix his equipment.
DX – “long distance” usually outside of the country of the broadcaster.
There is an entire set of Q-codes. These codes seek to have specific meanings and can be questions or answers – pretty handy when it comes to reducing typing!
- QRZ / QRZ? – “who is calling me.”
- QSB / QSB? – “you are fading / am I fading?”
- QRM / QRM? – “I’m suffering from interference / are you suffering from interference?”
- QRN / QRN? – “I’m suffering from noise / are you suffering from noise?”
- QRP / QRP? – “I’m running low power / are you running low power?”
- QSY / QSY? – “I’m changing frequency / shall I change Frequency?”
- QSL / QSL? – “I acknowledge receipt / do you acknowledge receipt?”
- QSO/QSO? – “I shall contact ____ / shall I contact ____ ?”
- and many more
Not surprisingly, the Q-codes have lost their verbish nature and become nouns as well. QSL becomes the acknowledgement and can have physical form in a QSL card, commonly sent in the mail to confirm a radio contact. QSO is the contact, not the process of making a contact.
The more difficult and time-consuming it is to send characters, the more one sees the use of ham abbreviations. My experience has been mostly with the PSK digital modes where characters are transmitted at roughly my typing speed, so extreme abbreviations are not really necessary. Inexperienced Morse code operators may be only able to punch out 5 characters/sec, so speaking in sentences is prohibitive. The RTTY digital mode also has its own form of rapid fire idiosyncratic exchange syntax that I have yet to master completely. The RTTY experts seem to relish making an exchange of call signs an information take no more than a few seconds and then moving on. Despite the common tendency for transferring only the most basic information very quickly, if you spend some time and practice a bit of verbal diplomacy, you can discover that there are real people with real personalities on the other end of the connection.
So for now,
TU for web QSO, 73 de AF7NX sk